Transformer coupling network



Feb. 16, 1954 R. c. OLESEN 2,669,697

TRANSFORMER COUPLING NETWORK Filed July 15, 1948 3 Sheets-Sheet 1 j 62 f0 2 z I 15 16 v 54 2 Z6 2 f2 44 ,0, 1 I! E 52 5 5 I 30 l 66 l g 3,24 11 1e Z2 i a! 0.5 no so 10 ax 0.5 m 50 10 INVENTOR:

RAYMOND C. 0L ESE/V WWW Feb. 16, 1954 c o' Es 2,669,697

TRANSFORMER COUPLING NETWORK Filed July 15, 1948 3 Sheets-Sheet 2 INVENTOR. RAYMOND C. OLESEN Patented Feb. 16, 1954 UNITED STATES PAT OFF I CE.

18 Glaims.

M'y-invention relates toelectrical signal repro ducing systems, and more particularly to. im provements in networks employing. transformers for coupling successive parts or stagesof such a-systemi It isconventional practice to employ atransformer to couple the input of an electric'ampli fier to the: output of another circuit; which for convenience is calledan input circuit. For. this purpose it is conventional to connect the primary winding of the transformer tozthe output -of'the input-I circuit and a secondary winding to the: input of the amplifier. In many application itis-desirable touarrange the connectionsinsuch azway'thatithe impedancewhich appears tobe connectedexternally across the. output of the input circuit is f-a constantresistanc value. For simplicity-the impedance which is-seen when measurements are made across a primary wind: ing of a transformer is. hereinafter referred, toas the virtual or apparent primary impedance, or sometimes simply as the primary impedance, or simply as the transformerinput impedance.

Itis conventional. to resort to impedance matching methods when-theinput circuitisdesigned topossessycertain knowncharacteristics when feedinginto a known constant resistance load", and theactual load to be supplied from the transformer, secondary hasanimpedance which doesnot equal the constant resistance mentioned: Such impedance matchingmethods are usedgior example,.when an electrical. amplifier having-a capacitive input is to b supplied. from aline-orotherinpu-t circuit which has an output impedance equal to the constant resistance mentioned..- Very often such output impedance is of low val'uesay about 400 to 6000 or lower, and veryoften when an amplifier is being fed from suchaline, a voltage step-up transformeris employed.-,

In: conventional impedance matching methods, it, istcustomary to connect a large resistance across the secondary winding of th transformer so :that the. virtual primary impedance. is 11011? reactive in character over a wide frequency range. The introduction of such a resistance across a. transformer causes a reduction in the gain that could otherwise. be obtained. with the transformer and, also causes an increase in the power consumed over the amount that would otherwise be consumed. This power consumption is generally minimized by employing atransformer of the power type. Even in the best circult designsof this type, the apparent input impedance of the transformervaries greatly with frequency over the range of frequencies of'interest.= Since such variation of the transformer input -impedance with frequency. causes undesim able frequency discrimination, this fa'ctz increases the 'dificulty' of calculating; in. advance the: frequency. characteristics that will be obtained: from any particular circuit designrin which an ampli fier isconnected'. to an input. circuit through a transformer.

The: object: of -1 my invention 1 is to: provide: an improved system of matching the input impedance of ,aztransformerawitlrthe outputiimped-arice of .an'input circuiti Accordingto my invention, impedance match ing is. accomplished. by employing atransformer coupling networlc which utilizes; a specially, designed equalizing circuit: connectedlto the-pri'-- mary winding of the transformer. The equalizing circuit includes anrim pedancewhich bears a reciprocal relationship with the virtual primaryimpedancevoi the transformer; Sinceimpedance matching may be accomplished: in-accord-ance with this invention without employing a resistor across theeseconclary winding, all the. disadvantages .attendantupon: theintroduction. ct I such a resistance-in an amplifier circuitareelimi nated and the use of voltage-type rather than power-type transformers for" impedance----matching iszfacilitateda The transformer coupling networks of my in:- ventionare. superiorto those previously eme ployedin' two important. respects. First,-v the characteristic; input-impedanceof my, type of transformer networlc is-more. nearly constant over any given range of: frequencies, thereby simplifying the design. of. amplifier. circuits. Secondly, with my transformer network it isposesible to. achieveeither a: higher amplification over a'predetermined 1 frequency band,. or corn versely, a predetermined amplification over a wider frequencyband. In either. of the. latter cases, the employmentofmy transformer coue' plingnetwcrk; results. in anincrease in the .sig nal-to-noise. ratio. at! the input of an electrical amplifier.useditherewith.

My; invention possesses manyother objects and advantages in addition to the foregoing, all of; whichwill be more readily apparent'from a consideration of exemplary embodiments of my invention. For this purpose four. formsofi transformer networks embodying the features ofmy invention are illustratedt'in' the" accompanying drawings which form a part'of thezpresentspeci fication. These forms of'"my"inventionrwillnow be described 'in" detail "in order to illustrate the general; Iprinciples'ofth'e invention: butiit is to be understcodthat the invention"isnctliinitedto the details of the forms illustrated and described, but that its scope is best ascertained by reference to the appended claims.

In the drawings wherein like legends in the several figures represent the same parts:

Figure 1 is a schematic wiring diagram of an electrical amplifier circuit employing one form of my transformer coupling network;

Figure 2 is a schematic wiring diagram representing the electrical characteristics of a transformer;

Figure 3 is a schematic wiring diagram of an equivalent circuit of the transformer impedance as viewed from the primary side thereof;

Figure 4 is a schematic wiring diagram of an impedance which bears a reciprocal relationship with the primary impedance of the transformer;

Figure 5 is a graph illustrating characteristics of my transformer coupling network;

Figure 6 is a schematic wiring diagram of another impedance which bears a reciprocal relationship with the primary impedance of the transformer;

Figure 7 is a schematic wiring diagram of an equivalent circuit approximating the primary impedance of the transformer;

Figure 8 is a schematic wiring diagram of an impedance which bears a reciprocal relationship to the approximate primary impedance of Figure '7 Figures 9 and 10 are schematic wiring diagrams of circuit elements which apply when a transformer supplies a generalized load;

Figure 11 is a schematic wiring diagram showing values of circuit elements employed in a particular coupling network;

Figure 12 is a schematic wiring diagram of a second form of my transformer coupling network;

Figure 13 is a schematic wiring diagram of a third form of my transformer coupling network; and

Figure 14 is a schematic wiring diagram of an electrical amplifier circuit employing a fourth form of my transformer coupling network.

FORM 1. GENERAL DESCRIPTION Referring to Figure 1, there is illustrated an electrical circuit in which a transformer coupling network l0 embodying features of my invention is employed to couple or interconnect an input circuit or line I2 with a load in the form of an audio-frequency amplifier M. More particularly, the input terminals l6 of the transformer coupling network Ill are connected to the respective output terminals [8 of the input circuit I2 and the output terminals 29 of the transformer coupling network are respectively connected to the input terminals 22 of the amplifier M. The coupling network ID of my invention has an input impedance which is a substantially constant resistance R0 in spite of the non-resistive character of the transformer 59 employed therein and the non-resistive character of the load supplied.

The electrical amplifier I4 comprises one or more electrical amplifying stages connected in tandem in amplifying relationship between the input terminals 22 and the output terminals 24. Only the amplifier tube 26 of the first or input stage is here illustrated. This amplifier tube 26 includes a control grid 28 and a cathode 30 which are respectively connected to the two input terminals 22. Preferably, the amplifier tube 26 is of a type and is operated under such conditions that it has a high input impedance, usually being operated for this purpose as a class A amplifier. In the practical applications with which my invention is primarily concerned, the input impedance is in the form of a grid-cathode capacitance C This capacitance includes two main parts, one part being due to the internal electrode capacitance existing between the grid 28 and the cathode 30 within the envelope of the amplifier tube 26, and the other part being due to the stray capacitance between the leads which connect the electrodes 28 and 32 to the input terminals 22 of the amplifier M. In practice the value of the capacitance Cg is usually made as low as possible in order to minimize attenuation of the amplifier at high frequencies. For example, in many well designed amplifiers employing input amplifier tubes 26 of the pentode type such as 6J7, the value of this input capacitance is usually of a low value of about 12 ,u F.

The input circuit i2 is here illustrated in the form of a cable which leads over considerable distance to a source 32 of electrical signals. The source 32 may be in the form of a microphone or a phonograph pick-up, or any other device which operates to translate vibrations into electrical signals which are to be amplified. The input circuit i2 may, however, be in the form of a filter network or may even be another amplifier. In the examples of the invention here considered, the input circuit 12 either possessess a characteristic impedance which is in the form of a substantially pure constant resistance R0 or is at least designed to feed into a circuit having a pure constant resistance R0 in order that the ini put circuit may possess certain specified characteristics over a predetermined band of frequencies. In an audio-frequency amplifier the band of waves to be amplified frequently extends from about 16 C. P. S. to 16,000 C. P. S.

The transformer coupling network ID of my invention which interconnects the input circuit 12 and the amplifier it is designed to have an input impedance (that is, impedance measured across the input terminals I6), which is a substantially pure constant R0 over the band of frequencies of interest. The transformer coupling network Ii! is also designed to amplify input voltages and to impress them upon the input of the amplifier M at a relatively high level with a minimum amount of attenuation of signals at high frequencies. To accomplish these results, an iron core vo'ltage-step-up transformer 50 and an equalizing circuit 50 designed in accordance with the principles of my invention are here employed.

The transformer 59 comprises a primary winding 52 and a secondary winding 5 wound upon an iron core 55. Opposite ends of the secondary winding 5 are connected across the output terminals 29 in order that output signals may be impressed upon the input terminals 22 of the amplifier i i. One resistor 82 of the equalizing circuit 66 is connected in series with the primary winding 52 across the input terminals l6 of the transformer coupling network I 0. Another resistor 6 and a reciprocal impedance network 66 of the equalizing network 60 are connected in series across the input terminal I6, thus being in parallel relationship with a circuit which includes the first mentioned resistor 62 and the primary 52.

In one way of practicing my invention the first resistor 62 has a value of resistance equal to the desired input resistance R0 diminished by the D. C. or low frequency resistance of the primary 51', winding-"52*, that is, a value'equal to R's-R5: and the'second resistor 64 has a valueof resistance equal to the desired input resistance Re. The impedance network 66- is so designed as to bear' a reciprocal relationship with the virtual prlmary' impedance of the transformer 50*, that is; the impedance as viewed looking into the pri mary winding 52 over the range of frequencies of signals to be amplified. More particularly; this reciprocal relationship is expressed by the equation I NR mi, Z1 zo(w) this equation w=21rf where f is: the frequency of any impressed" signal. Z'Mw') isthe'impedance at any frequency f; leokingintothe primary winding- 52 Audi also in this equation, Zr'fia) is the corresponding value of" the impedance of the network 66' at the samefrequency f. In this application, expressions such aSF(w) which are functions offrequen cy, are sometimes written as F('w)-, and some--- times simply written as F. In practice, the value of Zn; that is Zu(w-), is determined with a load connected across the output terminals which are equivalent to the input impedance (in this case the impedance of C of the amplifier to which the coupling network [0 is to be' connect'edi FORM 1. PRINGIPLES OF DESIGN OF TRANS-' FORMER COUPLING NETWORK.

The principles for designing a network such as the impedance network 66 which is the reciprocal of another known network are well known and are explained in various text books such as Communication Networks byErnst A. Guilleman (John Wiley and Sons, 1931). Before designing the impedance network 66, however, itis necessary to know the full nature of the input impedance Z0 of the transformer 56; An equivalent circuit of the transformer 50 and an equivalent of the virtual input impedance Z0 hich are applicable to an audio frequency amplifier M operated from a low impedanceline l2, are illustrated in Figures 2 and 3 respectively;

Referring to Figure 2, it is to be noted that the primary winding i2v of the transformer 50 may be represented by a series resistance Re, a leakage inductance L and an incremental primary inductance Lp all connected in series, and a resistance Re shunting the incremental inductance Lp. The resistance R5 represents the effective resistance corresponding to the power consumed by the core 56 due to eddy current losses. Also referring to Figure 2, it is to be noted. that the secondary winding 54 maybe represented by a secondary leakage inductance Ls, a secondary resistance Rs, and a secondary distributedcapacitance- Cs", all connectedin series across an incremental secondary inductance 115". In this equivalent circuit a coupling coefficient of unity exists between the incremental primary and secondary inductances Lip and Ls. The capacitance C9" of the load (not shown in Fig; 2 is connected in parallel with the secondary distributed capacitance CS. The equivalent circuit illustrated in Figure 2 neglects the capacitance existing between the primary and secondary windings 5'2 and 5 1, and the primary distributed capacitance for the reason that these capacitances are of negligiblevalue in many ap"-- plications. In the event that their values are not negligible, due account maybe takenof 'them Alsoin this equation- 6 byihose skilled in the artin accordance with the-principles of my inventionwithout" requir mg any further explanation here The input impedance Zo of the loaded trans former" which is represented by the equivalent circuit illustrated Figure 3; is derived from the circuit of Figure 2 by referring the various elementsof' the secondary winding of the latter to unity turns ratio. When this is done, the equivalent circuit includes an inductance 1'35, 9. Resistance Rs, and a capacitance Cs, all connected in series across the'incremental primary winding, inductance L and the parallel. resistance Re. The value ofthe inductancel'is, the resistance Re, and the condenser Cs are related to the constants of the primary and secondary windings 52- and. 54 and the input capacitance C3 of the amplifier M by the following equations:

where number of turns on secondary winding .l

number of turns on primary winding In Figure 4 thereis illustrated a network 66 which bears (a reciprocal relationship to the=*net'-x worksv illustrated in Figures 2 and 3 in accord ance with Equation 1. The network 66- comprises a resistance 1'1 and a condenser 01 connected in series with a parallel network 70 having three branches, which respectively corn-- prisea condenser 02, a resistance T2, and an in ductance 12. The values of the various circuit elements of the impedance network 66 are related to those of the equivalent network illustrated in Figures 2 and 3 and to the load by the following equations:

Where the value of any" factor suchasthe' pri= mary inductance Lp varies with signal strength, an average value applicable to the conditions of operation is employed. Since the resistance-r1- is actually in series with the resistance 64-, the two may be'replaced by a single resistancehaving a value equal to lit-+11: 4

7 Taking due account of the circuit relations expressed by Equations 2 and 9 inclusive, an analysis of the transformer coupling network l shows that the voltage E0 appearing at its output is related to the voltage E1 impressed upon its input from the line I 2 by the following equation:

(ROR.+Z (jwL,,+Z,)jwC,

and at median frequencies where R0 wZp Zs Re and Equation 10 reduces to Eom=nEi (14) Further analysis shows that the relative voltage amplification obtained with this circuit at low frequencies may be represented by the curve K1 illustrated in Figure 5. This curve K1 represents the equation E 1 ZE,- 3 (15) where The frequency ii of Equation 15 is the frequency at which the voltage amplification is down 3 db compared to the voltage amplification at median frequencies.

Likewise the relative voltage amplification at high frequencies is represented by a family of curves K2 of Figure 5. These curves represent the equation f f 1 1 fr 1% Q0 where f2, the series resonant frequency of the secondary winding, with the capacitive load attached is given by the equation l 18 f2 2 /L.c.

and the damping coeificient is:

f2 a Qo R0 R1 Graphs of the curves K2 representing the relative voltage amplification for several different values of damping coefficient Q0 about equal to unity are illustrated in Figure 5.

By virtue of the fact that the transformer coupling network l0 does not employ a resistor connected across the secondary winding 54 as in conventional practice as hereinabove explained, the transformer 50 may be of the voltage-amplifying type instead of the power-type. As a result,

higher gauge wire, that is, wire of smaller diameter, may be employed in the secondary Winding of the transformer 50 compared to the gauge of the wire required in impedance matching transformers of the power-type. By virtue of the fact that smaller-diameter wire may be employed in the secondary winding, all other things being equal, the secondary winding may be arranged in a group of mutually-spaced-apart sections, thereby resulting in a reduction of the value of the secondary distributed capacitance Cs and thereby resulting in an increase in the series resonant frequency f2 of the secondary winding. Conversely, more turns may be added to the secondary winding to attain a given series resonant frequency, thereby increasing the turns ratio 12. Thus the employment of such transformers in the coupling circuit I0 brings about the amplification of a wider band of signal frequencies for a given value of voltage amplification (or turns ratio). And conversely for a given frequency band, higher amplification at median frequencies may be obtained.

In practice the secondary winding is designed to yield a predetermined high value of resonant frequency ,fo near the top of the frequency band of signals which are to be amplified. The resonant frequency so obtained, which is given by Equation 18, is practically independent of the primary winding characteristics. Once the series resonant frequency f2 has been set, the turns ratio is then established at such a value as to yield the desired value of damping coefficient Qo given by Equation 19, taking due account of the value of the characteristic impedance R0 to be matched and the resistance of the secondary winding Rs referred to the primary. The turnsratio required for this purpose is determined very largely by the total leakage inductance Ls and the geometry of the core. The primary winding is then wound with the number of turns required to produce the desired leakage inductance L5 for the given geometry. The value of the primary incremental inductance Lp so obtained, determines the low frequency cut-off ii in accordance with Equation 16.

It is possible to obtain many of the advantages of my invention without departing from its fundamental principles by employing a resistance 62 having a value of resistance greater than R0Rc. In fact, for example, if the value of the resistance 62 equals Ru, the impedance network 66 should be replaced by one having an impedance given by the following relationship ii Z0(w) instead of relationship indicated by the Equation 1. The corresponding network is illustrated in Figure 6. This network is the same as the one Z1 formerly described, except that it is shunted by a resistor r3 given by the following equation:

Thus in general it appears that the damping coefficient Q which determines the shape of the amplification characteristics at high frequencies, .as .indicated by the family of curves K2, may be adjusted somewhat by increasing the value of .the resistance 62 and making compensation therefore by adjusting the value of the shunting .resistance m in the impedance network.

Expressed more broadly, if the'value of the resistance 62 is R and if the virtual primary impedance with any predetermined load connected across the secondary is Z000), the value of the reciprocal impedance 66 should be in order to achieve constant input impedance for the network it.

An approximate equivalent circuit for such a transformer 50 which is found to be satisfactory for many practical purposes is illustrated in Figure 7. For convenience, the impedance of this approximate electrical circuit is designated Za Comparing this circuit with that of Figure .3, it is to be noted that the sole difference lies in the fact that the resistance Rs has been moved from the load side to the input side of the iniductance Lp, thus being in series with the re- .sistor RC. Advantage may be taken of this equivalent circuit by setting the value of the resistance .52 equal to R0-Rc-Rs. In this case because .of the transfer of position of the resistance R5, the resistance r2 may be omitted .from'theimpedance network 66, thus forminga .network having a suitable reciprocal impedance Z .as illustrated in Figure 8. It is to be noted that the reciprocal of the practical equivalent circuit of Fig. .7 is given by the equation,

A transformer coupling network 10 employing the approximate impedance network Zp' is for vmany practical purposes as useful for impedance :matching "as the circuit arrangement hereinbe- :fore described employing the complete impedance network Z1 which is more nearly accurate :from a theoretical standpoint. In fact, the transformer coupling network based on the approximate equivalent circuit of Figure '7 has superior low frequency response compared to the one "based on the theoretical equivalent circuit "of Figure '3 because of the reduction in value of the resistance 62.

:It is to be'noted that even though the trans- ?former circuitnetworkmaybe arranged in differcent 'ways to obtain highly effective impedance matching, nevertheless in all of the cases ole-- scribed above, an equalizing circuit is employed which utilizes an impedance network which bears :a linear reciprocal relationship with the imped'ance looking into the transformer. In all cases however, the equalizing impedance network whichgives best results comprises a capacitance :c1 which bears a reciprocal relationship to the :primary incremental inductance Lp of the transformer connectedin series with a parallel resonance network which includes a capacitance cz and-an inductance Z2 which bear reciprocal relationships with the total leakage inductance Ls oof the transformer and the loaded secondary capacitance -CS respectively as viewedfrom the :primary circuit. These reciprocal relationships :have been pexpressedhereinbefore in Equations 5, 6, 7, and 9. It is to be noted in this connection 10 that the actual value required for the inductance Z2 depends not only upon the fixed value of the secondary distributed capacity CS, but also upon the capacity -Cg of the load which is to be matched, as explicitly set forth in Equation 4. Since, however, in many practical applications the input capacitance -Cg' of the load to be supplied is of about the same value which is small compared to the distributed capacity Cs, quite often in practiceit is perfectly feasible'to design the impedance network 66 in advance with the assurance that it will prove satisfactory in matching the input impedance of many ampli- .fiers l4. For this reason the input capacitance C of the amplifier It may be considered as .part of the impedance of the transformer 150.

On the other hand, if the transformer coupling network is to supply a different kind of load, this maybe taken into account by connecting the reciprocal of this load in series with the inductance 12 of the parallel network 10. In fact, if the coupling network I0 is to feed into a more generalized load Zr, the capacitance C5 of the equivalent circuits should be replaced by an impedance ZS given by the following equation,

where CS is the distributed secondary capacity of the transformer. This impedance is represented in Figure 9. Account may be taken of themore generalized load by Connecting the ciprocal Zr. of the generalized load in series with the parallel inductance Z2 of the impedance network as illustrated in Figure 10. The parallel network "l0 thus has one branch comprising an impedance Z in which the reciprocal Z2 of the generalized load is connected in series with the reciprocal Z2 of the secondary distributed capacity. The value ofZz is given .by theequation Z2 ZL and the --value of 12 by the equation 12:(7LRO) 0s (26) In any event, by following the principles of design outlined, a transformer coupling network having a substantially constant input resistance R0 may be constructed and thus .improved impedance matching obtained.

.Form 1.E:cample In a particular application in which a transformer coupling network Ii) was designed to match a GOO-ohm line 12 with an amplifier 14 .having an input capacitance .of C equal to 12 F, a shell-type transformer 50 was employed having the following characteristics:

Number of turns on primary The corresponding .set :of circuit constants sQf the impedance network employed therewith ll in order to match a GOO-ohm line were as follows:

C1 40 pf.

l2 3.70 mh.

It is to be noted that the approximate equivalent circuit of Figure 7 and the corresponding reciprocal impedance of Figure 8 were employed in this example. The corresponding equivalent of the coupling network I is illustrated in Figure 11. In this equivalent network the input capacitance of the amplifier I4 is treated as part of the network and an ideal transformer 80 having perfect coupling and a turns ratio n and an input impedance large compared to Cs, is represented in the output.

Tests of the transformer coupling network employing the particular transformer mentioned and the corresponding impedance network showed that, in fact, the apparent input resistance measured across the input terminals I6 of the coupling circuit III was nearly constant over the band of frequencies to be amplified. The actual measured values of resistance at difierent frequency are indicated in column A of Table I.

In column B of this table, corresponding values are given for the input resistance when the series capacitance c1 of the equalizing circuit was omitted as shown in Fig. 4. And in column C similar data is given for a power-type transformer connected in conventional manner without employing any equalizing circuit. The superiority of the results to be obtained by employing an equalizing circuit are obvious from the table. In particular, it is to be noted that improved results are obtained at high frequencies, both with and without a series condenser 01 (see Fig. 4), and that good results are obtained at low frequencies too when the series condenser 01 is employed.

Form 2 In Figure 12 there is illustrated a second form of impedance matching network III] which employs a transformer I56 which has the same characteristics as that described hereinabove but which has a different equalizing circuit I66. This equalizing circuit comprises a resistor I62 which shunts the primary winding I52 of the transformer I50, and also comprises a parallel network I63 which includes a resistance I64 and a reciprocal impedance network I 66. The values of the two resistances I62 and I64 are each equal to R0, that is, the value of the constant resistance impedance desired across the input terminals II6. In this particular case, since the internal resistance R0 of the primary winding I52 is not lumped with some other resistance, an impedance network I66 having an impedance Zz' of the type illustrated in Figure 6 is employed, The

coupling network III] possesses the same overall characteristics as the one formerly described, the ratio of the output voltage E0 appearing across the output terminals I20 bearing the same relationship to the voltage Er impressed upon the input terminals II6 as expressed by Equations 10 to 19 inclusive, taking due account of the inclusion of the resistance R0 in series with the primary winding and the shunting resistance n of the impedance network I66.

Because of the manner in which the primary winding resistance Re appears in this circuit I III, the overall characteristics are the same as those of the variation of the first form II) employing the impedance network Z2 and is therefore not quite as satisfactory as the variation of the first form I0 employing the impedance network Z1 because of the resultant loss of low frequency response.

Form 3 A third form of transformer coupling network 2IIJ illustrated in Figure 13 is particularly suitable for supplying signals to an amplifier 2I4 having a balanced input represented by three input terminals 222. This transformer coupling network 2 It employs three corresponding output terminals 220 which are respectively connected to the three input terminals 2 III of the amplifier 2I4 but employs only two input terminals 2I6 adapted for connection to a corresponding input circuit. The coupling network 2I0 employs two transformers 250 each having a corresponding primary winding 252 and a secondary winding 254. The respective secondary windings 254 are connected between adjacent output terminals 226 of the coupling network 2 I0. Each of the transformers has the same impedance Z0 as viewed from the primary, and for this purpose two transformers 250 may be employed identical in characteristics with the first transformer 50 described above.

In this network 2I0 an equalizing circuit 260 is employed which includes a single resistance 263 and two like impedance networks 266. These elements and the primary windings 252 of the transformers 250 are arranged in a lattice or bridge network. When considered as a lattice network, the two primary windings 252 are connected in opposite sides thereof in series with the load resistance 263 and the two impedance networks 266 are diagonally connected to the load resistance 263 from opposite sides of the input. When considered as a bridge it is to be noted that the two primary windings 252 are arranged in one pair of coniugate arms and the two impedance networks 266 are arranged in the remaining pair of conjugate arms and the resistance 263 is connected in the diagonal or output position.

When the resistance 263 has a value equal to R0 and each of the impedance networks 266 is the reciprocal of the transformer input impedance Z0, the output voltage En appearing across the secondary winding 254 of each transformer is equal to the output voltage appearing across the secondary winding of the transformer of either of the first and second transformer networks I0 and III). Thus it appears that even when the input resistance of the third form of network 2I0 is the same as that of the first and second forms transformer networks described, the total voltage appearing across the extreme output terminals 2H1 is I WIGQ that obtained with the first two forms,

inseam? 13 Form 4 .Ifhe fourth form of transformer coupling network .3l0 illustrated in ,Figure 14 is somewhat similar to the third form illustrated in Figure 11,

but employs only a single transformer 3'50 and only two output terminals 320. Tiro -transformer 350 employs a single secondary' winding 354 conv.nected between the output terminals .320, but also employs two primary windingsBfiZ whichare connected in opposite sides of the lattice network 360. These connections of the primary windings 352 in the lattice networkare the same as those described in connection with the third form of coupling vnetworkZH'l, the only difference being in the fact that the two primary windings 352 .arenow mounted upon a common. coreSEB. The two primary windings 352 are mutually coupled in one-to-one relationship and carry ec ual .currents due to their symmetrical positions in the network. The twoprimary windings 3'52 are con- ,nected so that the voltages induced in the secondary winding 354 are in phase.

Because of this coupling of the two primary windings 352, the number of turns in each primary winding is made 0.70 of the number employed in the primary windings of the transfformers previously described herein, in order to ,produce the same impedance Z0 looking into each of'the primary windings. This relationship applies as long as the primary windings 352 are operated simultaneously in a symmetrical, network. Due to the reduction in the number of turns in each of the primary windings, the turns ,ratio of the secondary winding with respect to "each of the primary windings isno longer n .as "formerly, but is now 1.411.. This means that for a given secondary winding the voltage appearing .acrossthe output terminals 3'2ilisnow 1.4 min- .stead of E0. In other words, by splitting the .pri-

mary winding of a transformer and connecting it'in a lattice network as describe'd,and increas ing the total number of turnson theprimaryby a factor 1.4, a larger output signal is obtained than in Forms 1 and 2, but a smalleroutput than in Form'3.

CONCLUSION From the foregoingdescription of the several :forms of my invention it is clear that I have provided an improved transformer couplingnetwork :which possesses a substantially constant resistance over a wide frequency range, and which is particularly suitable for matching the constant resistance output impedance of some signal source with the input impedance of an amplifier. cording to my invention such impedance matching is obtained without introducing undesirable and unexpected discrimination because of differ- :ences of transformer inputimpedances .at different frequencies. Furthermore, fromtheforegoing-description it will'be appreciated that :my .improved transformer coupling network is capable of producing higher signal gain over .a predetermined frequency band than is obtainable with a conventionalimpedance matching circuit, and that as a result my circuit may be employed for improving the signal-to-noise ratio at the input of an amplifier which is supplied by a trans- .former.

Though my invention has been described-with particular reference to audio amplification systems employing iron-core transformers, .it is :egually applicableto other systems,xsuchrasivideo amplification systems. employing :ainmore trans- :formers. w

"Winnie :my invention has been described more or less in detail with reference to centaimembodiments thereof, it will be understood that many changes. may hemade therein without departing from the principles of myinvention. jIt istherefore to be understood that many modifications and changes may be made in the details of the circuits described, as will now appear to those skilled .in the art, without departing from the -nature of my invention but still within the scope ..of the appended claims.

I .claim:

.1. Juan alternatingcurrent circuit having an input and an output and adapted .to transmit alternating currents over a predetermined range of frequencies from .said input to said output, a transformer having a primary winding and a secondary v winding, said secondary winding being connected'to said output, said secondary winding resonating atafrequency in,saidfreguency-range, .said transformer having an impedance Zo(-w-) when viewed from said input, -a compensating resistance-meanshaving airesistance value offl :and 1a 'reciprocal network having an impedance .Z'(w) related to said impedance Zone) :by the requation I R0 Z 00") said compensating resistance means and :said reciprocal network being-so connected between said input and said-primary windingas to render the impedancemeasured across said input'a substantially constant resistance over said frequency range.

.2. An alternating current circuit as defined in .claim 1 wherein said impedance :Zo comprises the reactance of an inductance Lsand the :re- ,actance of capacitance Cs connected in -series :and said reciprocal network comprises .a compensating inductance Z2 and a compensating capacitance ('02 connected in parallel, the :.value .of .said..compensatin g inductance beinggivenbyzthe equation.

and the Val-1620f said capacitance being given bythe equation 3. In an alternating current circuithaving an input and an output, a transformer having -a primary winding and a secondary winding, said secondary windin being connected to said output, a resistor R0 connected in series with said primary winding across said input, said trans- ..for mer having animpedancezow) .when viewed from said .input, {and a compensating resistance .30 and ,a compensating .-network connected in series across said input, said compensat ns'network having an impedance Z'(w) related to said impedance Zo(w) by the equation "whereby the impedance measured: across said input 'is a-substantially constant resistance over --a mangeof frequencies of alternating currents applied thereto.

:4. .An alternating current circuitas'defined in claim 13 wherein said impedance Zena) comprises -;the:reactance of an inductance'Lsand the reactance ,.of capacitance .Cs connected in series and .zsaidzcompiensating:network comprises: an "induct- 81106 Zaiaud a roapacitance F02 connected "in mml5 allel, the value of said inductance being given by the equation Z2=R0 Cs and the value of said capacitance being given by the equation 5. An alternating current circuit as defined in claim 4 wherein said impedance comprises a resistance Re that is connected across said inductance L5 and said capacitance Cs and wherein said compensating network comprises a resistor r1 given by the equation connected between said compensating resistance R0 and said inductance and capacitance.

6. An alternating current circuit as defined in claim 5 wherein said impedance comprises an inductance Lp that is also connected across said inductance L5 and said capacitance Cs.

7. An alternating current circuit as defined in claim 6 wherein said compensating network also comprises a condenser 01 given by the equation connected in series with said resistor T1 between said compensating resistor R0 and said compensation inductance Z2 and capacitance c2- 8. In a transformer network having an input and an output a transformer having a primary winding and a secondary winding, said secondary winding being connected acros said output, the equivalent circuit of the virtual primary impedance comprising an incremental primary inductance and a series circuit connected across said incremental inductance, said series circuit being represented by a total leakage inductance Ls connected in series with a capacitance n CS, where C5 is the secondary distributed capacitance of said transformer and number of turns on secondary winding number of turns on primary winding and a parallel network having a compensating capacitance and a compensating inductance connected in parallel therein, said compensating capacitance 02 having a value that is the reciprocal of said leakage inductance, which value is given by the equation and said compensating inductance 12 having a value which is approximately the reciprocal of the secondary distributed capacitance as viewed from the primary winding and which is given approximately by the equation Z2: (nRo) Cs' ductance and a -series'circuit connected across number of turns on secondary winding number of turns on primary winding a compensating capacitance c2 having a value that is the reciprocal of said leakage inductance, which value is given by the equation and a compensating inductance 12 having a value which is approximately the reciprocal of the secondary distributed capacitance as viewed from th primary winding and which is given approximately by the equation 12: (1L Ru Cs means connecting said compensating capacitance and said compensating inductance in parallel in a parallel network, said parallel network and said primary winding being connected in parallel, and resistance means having a value R0, said resistance means, said compensating capacitance and said compensating inductance being so connected and their values being such that said transformer network possesses a substantially constant resistance R0 over said frequency range.

10. In a transformer network having an input and an output, the combination which comprises: a transformer having a primary Winding and a secondary winding, said secondary winding being connected across said output, said transformer as viewed from the primary winding being representable by an equivalent circuit which includes a leakage inductance Ls connected in series with a loading capacitance Cs whereby the virtual impedance measured across the primary winding is a variable function over a predetermined frequency range including the frequency at which said leakage inductance and said loading capacitance resonate, and an equalizing circuit comprising a network including a compensating capacitance c2 and a compensating inductance Z2 connected in parallel therein, said compensating capacitance and said compensating inductance being characterized by having values that are related to said constant resistance, said leakage inductance, and said loading capacitance by the following equations:

said parallel network and said primary winding being connected in parallel, and resistance means having a value R0, said resistance means, said compensating capacitance and said compensating inductance being so connected and their values being such that said transformer network possesses a substantially constant resistance R0 over said frequency range.

11. In a transformer network having an input and an output for transmitting from said input to said output audio frequency waves in a predetermined frequency range, and including a transformer having a primary winding and a secondary winding wound on an iron core, said secondary winding being connected across said output, the equivalent circuit of the virtual primary impedance comprising an incremental primary inductance and a series circuit connected across said incremental inductance, said series circuit being represented by a total leakage inductance Ls connected in series with a loading capacitance Cs, and equalizing means including resistance means and a parallel network connected between said primary winding and said input, said parz-z-ilel network having a compensating capacitanc C2 and a compensating inductance 22 connected in parallel branches thereof, said capacitance 02 having a value that is the reciprocal of said leakage inductance and which is given by the equation and said inductance Z2 having a value that is the reciprocal of said loading capacitance and which i given by the equation where the resonant frequency Of said series circuit and the anti-resonant frequency of said parallel network are equal and lie in said frequency range, said resistance means being so connected and proportioned that the impedance measured across said input is a substantially constant resistance R0, whereby amplitude distortion caused by said leakage inductance and said capacitance is compensated in said frequency range.

12. In a transformer network having an input and an output for transmitting from said input to said output audio frequency waves in a predetermined frequency range, and including a transformer having a primary winding and a secondary winding wound on an iron core, said secondary winding being connected across said output, the equivalent circuit of th virtual primary impedance comprising an incremental primary inductance and a series circuit connected across said incremental inductance, said series circuit being represented by a total leakage inductance Ls connected in series with a loading capacitance Cs, means connecting said primary winding across said input, and an equalizing means including resistance means and a parallel network connected across said input and across said primary winding, said parallel network having a compensating capacitance c2 and a compensating inductance 12 connected in parallel branches thereof, said compensating capacitance 02 having a value that is the reciprocal of said leakage inductance and which is given by the equation and said compensating inductance 12 having a value that is the reciprocal of said loading capacitance and which is given by the equation where the resonant frequency of said series circuit and the anti-resonant frequency of said parallel network are equal and lie in said frequency range, said resistance means being so connected and proportioned that the impedance measured across said input is a substantially constant resistance R0, whereby amplitude distortion caused by said leakage inductance and said capacitance is compensated in said frequency range.

13. In a transformer network having an input and an output for transmitting from said input to said output audio frequency waves in a predetermined frequency range, and including a transformer having a primary winding and a secondary winding wound on an iron core, said secondary winding being connected across said output, the equivalent circuit of the virtual primary impedance comprising an incremental primary inductance and a series circuit connected across said incremental inductance, said series circuit being represented by a total leakage inductance Ls connected in series with a loading capacitance Cs, and equalizing means connected between said primary winding and said input, said equalizing means comprising a first resistor having a value Ru connected across said primary winding, said first resistor and said primary winding forming a first parallel circuit, said equalizing circuit also comprising a second parallel circuit connected in series with said first parallel circuit across said input, said second parallel circuit comprising two parallel branches, one of said branches comprising a resistor having a valu R0 and the other of said branches comprising a parallel network having a compensating capacitance c2 and a compensating inductance 12 connected in parallel branches thereof, said compensating capacitance 02 having a value that is the reciprocal of said leakage inductance and which is given by the equation and said compensating inductance Z2 having a value that is the reciprocal of said loading capacitance and which is given by the equation where the resonant frequency of said series circuit and the anti-resonant frequency of said parallel network are equal and lie in said frequency range, whereby amplitude distortion caused by said leakage inductance and said capacitance is compensated in said frequency range.

14. In a transformer network having an input and an output for transmitting from said input to said output audio frequency waves in a predetermined frequency range, and including a secondary winding means connected across said output, a pair of primary windings inductively coupled by iron core means to said secondary winding means, the equivalent circuit of the virtual primary impedance of each primary winding being the same and comprising an incremental primary inductance and a series circuit connected across said incremental inductance, said series circuit being represented by a total leakage inductance LS connected'in series with a loading capacitance C5, the improvement which comprises: an equalizing circuit operatively connecting said primary windings to said input for rendering the impedance measured across said input a substantially constant resistance R0 over said frequency range, said equalizing circuit com prising a resistor having a value equal to said constant resistance Ru connected between two ends of the respective primary windings, the remaining ends of said primary windings being connected to opposite sides of said input and also a pair of parallel networks diagonally connecting the respective sides of said input to the respective two ends of said primary windings, each of said networks comprising a compensating capacitance C2 and a compensating inductance Z2 operatively connected in parallel branches thereof, the compensating capacitance oz of each parallel network having a value that is the reciprocal of said leakmosa cs? 5119 age inductance and which. :is; givensby the :equation -.,circuit and theanti-resonant frequency of. .each :said parallel networkare equal and lie in said frecaused by said leakage inductance and thesaid -,capacitance is ..compensated .in said frequency range.

.15.A transformer. network as defined in. claim ssured-iacross-the primarywindingbeingm-variable function ;Zo(w) of Y frequency throughout :33 ,1 irequency: range overlapping a: resonantrfrequenoy, "the damping coefiicientt oi the secondary :winding at. said resonant frequency being aboutsunity, am equalizing network-comprising the combination of: a; firstsresistor rconnected: insseries- 'withrsaid primary windingzacross-said input, the valuezof -said first-resistor :being R=R0-.Rc :.-.where;Rc-,-is

:10 the e'fiective seriesrresistance: of said transformer viewed from the primary: side'xthereof, awsecond .resistor and;;a.-'compensating network, having i3 reciprocal impedance Z (w) said second;resistor and said compensating network being connected 'quency whereby amplitude ns in series with each othercacross said input, said second resistor having a value R0, said reciprocal -.impedance Z'(w) bearing a linearrreciprocalre- :lationship to -,a circuit comprising. said: firstinesistor and said virtual impedance .zuwl lt .Mwherein said. magnetic coremeans,comprises whereby the impedance looking into the input of a. single magnetic core inductivelylinking said secondary windingv means. and both of said primaryiwindings.

16.;In a transformer. network having. an input said transformer network is a substantially constant resistance having a value R0.

18. Ina transformer network having .an input and an output andlincluding a transformer havand an output, the combination which comprises: {mg a: primary Winding and a Secondary winding,

a. transformer having a primary winding. .andla ,secondarywinding, said secondary windingibeing connected across said output, said transformeras viewed from the primary winding being represaid secondary windingbeing connected across said ,output, thevirtualimpedance Z0(w) mea- ,sured across -;.the primary winding being a variable "function "Zo(w) of frequency, the dam n sentable by an equivalentcircuit which includes 530 coefficientptthasecondary winidmgv atresonanc'e a leakage inductance Ls connected in series with ,a loadingcapacitance Cs whereby the virtual impedancelmeasured acrossthe, primary winding is a variable function over a predetermined fre- .,being..a out un ty a qu network com- .prlsing thecombinationpf a first resistor having a valuetof R connectedrin eries with said ,primary wmding across said input, av second.- requency range including the"frequency at which 1- sistol. pensating network havinga resaid leakage inductanceand said loading capacitance resonate, and an equalizing circuit interconnecting said primary winding and .said' input, said equalizing :circuit comprising resistance ,ciprocal impedance Z'(w) connected in series with each other ,across said'input, said second resistor havinga 'valueiBo, and said impedance Z' (w) I being related .to-.sai.d virtual,impedance means an'da parallel network including a comflbyh-thewequation -pensating capacitancecz and a compensating inductancelz connected in parallel therein, said compensating capacitance and said compensating .inductance'being characterized byivalues that are related to said constant resistance, said-leakage followin equations:

iRu

lsaid resistance-means,said. compensatingzcapacistance and said compensatingx inductance beingso connected; and their .values ,beingsuch ,that said -having a primarywinding-and asecondarywind- 1-160 -ing, said secondary winding being connected across? said output, the virtualrimpedance; meainductance, and said loading "capacitance by the o Z6((J)1+IRQRQ whereby the impedance looking into the input of said transformernetwork is a substantially :ccnstant resistance having a value R0.

RAYMOND C. OLESEN.

*References: Cited in the file of this patent 

